RF system using AM with orthogonal offset

ABSTRACT

An RF system using amplitude modulation (AM) with orthogonal offset is disclosed. The orthogonal offset generator can shift the AM signal trajectory away from the origin while maintaining the time domain requirements for an RFID signal, such as waveform edge rise and fall times. In some embodiments stored waveforms incorporating the controlled orthogonal offset are used to synthesize a sequence of symbols. The stored waveforms may also include nonlinear and/or linear predistortion to reduce computational complexity. The waveforms can be represented in Cartesian coordinates for use in a direct conversion transmitter or polar coordinates for use in a polar modulation transmitter. An RFID system can also include a receiver to receive incoming RFID signals.

CROSS-REFERENCE TO RELATED APPLICATIONS

This application is a continuation-in-part of and claims priority fromcommonly owned U.S. patent application Ser. No. 15/711,458, which is adivisional application of and claims priority from commonly owned U.S.patent application Ser. No. 15/110,994, now U.S. Pat. No. 9,813,115,which is a national phase application of and claims priority fromcommonly owned PCT Patent Application Serial Number PCT/US2014/068484.The entire disclosure of all of these prior applications is herebyincorporated herein by reference.

BACKGROUND

The Gen2 RFID protocol includes an amplitude modulation (AM) mode,referred to as phase reversal amplitude shift keying (PR-ASK) in theprotocol specification, which can achieve good spectral occupancy.However, the amplitude of the radio frequency (RF) signal passes throughzero. This gives 100% amplitude modulation depth and produces phasediscontinuities with 180 degree phase jumps. Many RFID readers aredesigned using RF power amplifiers (RFPAs) with high power efficiency,such as class-AB or class-C. These high efficiency power amplifiers tendto have reduced linearity as compared to class-A amplifiers. These highefficiency power amplifiers work better when the amplitude modulation ofthe transmission signal is reduced. While ideal PR-ASK modulation canachieve good spectral occupancy, the 100% AM depth and the phasediscontinuities can create significant distortion for power efficientRFPAs commonly used in RFID readers. This distortion causes spectralregrowth which can significantly degrade the spectral occupancy.

Some RFID reader designs include nonlinear predistortion on the basebandtransmission signal to improve overall transmitter linearity andmitigate the spectral regrowth problem. This nonlinear predistortion iscalculated using input-output characterization of the RFPA. Theinput-output characterization can be very difficult to achieveaccurately when the transmission signal modulation depth is at or near100%, such as with PR-ASK. If digital predistortion is used, theamplitude and phase distortion of the RF power amplifier is difficult tomeasure when the RF signal trajectory goes through the origin. Thus,ideal PR-ASK has good spectral occupancy but is difficult for many powerefficient RFPAs to reproduce linearly and furthermore is difficult toimplement RFPA predistortion for due to its deep amplitude modulationdepth.

SUMMARY

Embodiments of the present invention provide apparatus and methods forgenerating new forms of amplitude modulation (AM) transmission signalsin a radio frequency (RF) system. The apparatus according to embodimentsof the invention can for example be used in an RFID system to generatetransmit signals fully compliant with the ISO-18000-63 or EPCGlobal Gen2specification and having spectral occupancy better than conventionalPR-ASK signals, but without the fully modulated amplitude wherein thesignal trajectory passes through the origin. The transmit signalsynthesis can support both direct conversion radio architecture andpolar modulation architectures.

To improve transmit data rates, some embodiments of the presentinvention use M-ary AM to send commands to the tags. In M-ary AM, datais encoded using a set of “symbols” comprising multiple amplitudevalues. Typically we would use M=2^(m), m being an integer. Thus, m bitsare encoded in each symbol period, thereby increasing transmit datarates. More generally some angle modulation could be included, but thisdisclosure will universally refer to the reader's transmit signal as anAM signal, since current passive tags can only decode amplitudemodulation, they cannot decode angle modulation. An RF system accordingto at least some embodiments of the invention includes an AM signalgenerator together with an orthogonal offset bias generator connected tothe AM signal generator output. This AM transmit signal may have deepmodulation depth, near 100% or fully modulated. The orthogonal offsetgenerator can shift the transmit signal trajectory away from the originto create a type of amplitude and phase modulated signal which hascontinuous phase modulation and reduced amplitude modulation depth as aresult of a controlled orthogonal offset. The resulting offset shiftedtransmit signal is substantially easier for many RFPAs to reproduce withgood quality due to this reduced AM depth. Orthogonally offset transmitmodulation will be denoted “OAM” herein for offset-AM.

An RF system according to at least some embodiments of the inventionincludes a PR-ASK signal generator together with an orthogonal offsetgenerator connected to the PR-ASK signal generator. The PR-ASK signalgenerator can produce a fully modulated AM signal representing a symboland/or a sequence of symbols. The orthogonal offset generator is capableof shifting the PR-ASK signal trajectory away from the origin to createanother type of amplitude and phase modulated signal which hascontinuous phase modulation and reduced amplitude modulation depth as aresult of a controlled orthogonal offset. The resulting offset shiftedPR-ASK signal is substantially easier for many RFPAs to reproduce withgood quality due to this reduced AM depth. Orthogonally offset PR-ASKmodulation will be denoted OPR-ASK herein.

In an RFID system, the resulting offset shifted signal maintains thetime domain requirements for an RFID waveform, such as edge rise andfall times and the symbol edge-to-edge timing specifications. Theresulting offset shifted signal maintains the frequency domainrequirements for an RFID waveform with improved spectral occupancy. TheRF system can also include an RF source to produce a carrier wave, andan RF amplifier connected to the RF source and the orthogonal offsetgenerator for transmitting the sequence of symbols modulated onto thecarrier wave to produce an AM signal as the transmitter waveform.

In other embodiments of the invention, the RF system may convert asignal from Cartesian to polar representation. Unlike PR-ASK which hasphase discontinuities of 180 degrees and is extremely difficult togenerate with a polar transmitter, the OAM modulation has continuousphase modulation which is readily produced in a polar transmitter. Thepolar representation can be used as input to a polar modulationtransmitter system, thus producing a sequence of symbols modulated ontothe carrier wave as a polar signal.

In some embodiments of the invention, the RF system can also include adigital predistortion block connected after the orthogonal offsetgenerator. In some embodiments of the invention, a transmitter nullingoffset generator can be included to cancel nuisance offsets introducedin the transmit baseband circuitry and RF mixer. In some embodiments thetransmitter nulling offset generator can be implemented at least in partby a summer. In some embodiments of the invention, the RF systemincludes a gain and phase imbalance equalizer to cancel nuisance gainand phase mismatch between the in-phase and quadrature-phase paths ofthe transmit baseband circuitry and RF mixer.

In at least some embodiments of the invention, stored waveforms, whichmay incorporate the orthogonal offset, are used to synthesize a requiredsequence of RFID symbols as a transmitter signal. Each stored waveformrepresents an RFID symbol. The stored waveforms may also includenonlinear and/or linear predistortion applied to substantiallyundistorted waveforms to reduce computational complexity in the digitalsignal processor. The stored waveforms may also mathematically include arotation about the origin. In some embodiments the waveforms arerepresented in Cartesian coordinates for use in a direct conversiontransmitter. In some embodiments, the stored waveforms are representedin polar coordinates for use in a polar modulation transmitter. In anyembodiment the RFID system can include a receiver to receive incomingRFID signals.

The waveforms can be stored in a storage medium such as a memory and caninclude a reference waveform and a reversed phase version of thereference waveform for each symbol. A multiplexer can be connected tothe storage medium to select one of the waveforms in accordance with thesymbol required in the sequence at a given time. Two multiplexers and aphase select switch can alternatively be used.

In operation, a processor repeatedly determines a current polarity statefor a stored waveform corresponding to an RFID symbol in a sequence ofRFID symbols, where each waveform is an OAM waveform that includes anorthogonal offset as described above. The processor retrieves eachwaveform from a storage medium, either a reference waveform or areversed phase version of the reference waveform in accordance with thecurrent polarity state. Each reference waveform represents an RFIDsymbol usable in the sequence of RFID symbols. The processor uses thesewaveforms to assemble the sequence of RFID symbols. The processor,together with a computer usable storage medium, such as a memory tostore waveforms and executable computer program code or firmware, can beused as the means to carry out a transmitter synthesis embodiment of theinvention.

BRIEF DESCRIPTION OF THE DRAWINGS

FIG. 1 is a functional block diagram of an example operating environmentfor an RFID system with an amplitude modulated reader-to-tagcommunications link.

FIG. 2 is a block diagram of an RFID reader using a direct conversiontransmitter for generating the amplitude and phase modulatedreader-to-tag communications link.

FIGS. 3A, 3B, and 3C are plots of a PR-ASK signal as may be used in anRFID reader transmission.

FIG. 4 is a phase plane graph to illustrate the signal trajectory ofPR-ASK.

FIGS. 5A, 5B, and 5C are plots of the newly disclosed OPR-ASK signal asmay be used in an RFID reader transmission.

FIG. 6 is a phase plane graph to illustrate the signal trajectory ofOPR-ASK.

FIG. 7 is a block diagram of an embodiment for generating the OPR-ASKtransmit signal within a DSP in a direct conversion RFID reader system.

FIG. 8 is a block diagram of an alternative embodiment for generatingthe OPR-ASK transmit signal within a DSP for a polar modulation RFIDreader system.

FIG. 9 is a block diagram of a transmit signal synthesis technique usingstored waveform tables.

FIGS. 10A, 10B, 10C, and 10D are plots showing example OPR-ASK waveformsas would be used in the stored waveform table based synthesis techniqueof FIG. 9. FIGS. 10A-10D provide an example for direct conversionradios.

FIGS. 11A, 11B, 11C, and 11D are plots showing example OPR-ASK waveformsas would be used in the stored waveform table based synthesis techniqueof FIG. 9. FIGS. 11A-11D provide an example for polar modulation radios.

FIG. 12 is a block diagram of an alternative transmit signal synthesistechnique using stored waveform tables.

FIG. 13 is a flowchart illustrating the process for creating OPR-ASKwaveforms, whether for storage in a waveform memory tables or for outputto a transmitter.

FIG. 14 is a flowchart illustrating the process for synthesizing OPR-ASKtransmit signals to communication with RFID tags in an RFID system.

FIGS. 15A and 15B are plots of an AM transmit signal as may be used inan RFID reader transmission.

FIGS. 16A and 16B are plots of the newly disclosed OAM signal as may beused in an RFID reader transmission.

FIG. 17 is a block diagram of an embodiment for generating the OAMtransmit signal within a DSP in a direct conversion RFID reader system.

FIG. 18 is a block diagram of an alternative embodiment for generatingthe OAM transmit signal within a DSP for a polar modulation RFID readersystem.

FIG. 19 is a flowchart illustrating the process for creating OAMwaveforms, whether for storage in a waveform memory tables or for outputto a transmitter.

DETAILED DESCRIPTION

Embodiments of the present invention now will be described more fullyhereinafter with reference to the accompanying drawings, in whichembodiments of the invention are shown. This invention may, however, beembodied in many different forms and should not be construed as limitedto the embodiments set forth herein. Like numbers refer to like elementsthroughout.

Terminology used herein is for the purpose of describing particularembodiments only and is not intended to be limiting of the invention. Asused herein, the singular forms “a”, “an” and “the” are intended toinclude the plural forms as well, unless the context clearly indicatesotherwise. It will be further understood that the terms “comprises” or“comprising,” when used in this specification, specify the presence ofstated features, steps, operations, elements, or components, but do notpreclude the presence or addition of one or more other features, steps,operations, elements, components, or groups thereof. The term “block”can be used to refer to hardware, software, or a combination of the twothat performs a particular function, group of functions, step, orcollection of steps. Comparative, quantitative terms such as “less” or“more”, are intended to encompass the concept of equality, thus, “less”can mean not only “less” in the strictest mathematical sense, but also,“less than or equal to.”

Unless otherwise defined, all terms (including technical and scientificterms) used herein have the same meaning as commonly understood by oneof ordinary skill in the art to which this invention belongs. It will befurther understood that terms used herein should be interpreted ashaving a meaning that is consistent with their meaning in the context ofthis specification and the relevant art and will not be interpreted inan idealized or overly formal sense unless expressly so defined herein.It will also be understood that when an element is referred to as being“connected” or “coupled” to another element, it can be directlyconnected or coupled to the other element or intervening elements may bepresent.

This disclosure has to do with generating signals for RFID transmission.The conventional approach to generating modulated transmission signalsis to use digital signal processing (DSP) techniques which produce a“digital signal”, which is a quantized and sampled signal. These digitalsignals may be denoted such as x(n), where the argument is a variablesuch as n, m, or k. These variables represent the sample index,typically under a uniform sampling period Ts. The digital signal ispassed from the DSP to a digital-to-analog converter (DAC) whichproduces a continuous time version of the signal, commonly using azero-order-hold representation at the DAC output. For band limitedsignal reproduction the DAC output is followed by a low pass or bandpass reconstruction filter which produces a continuous time, continuousamplitude version x(t) of the digital signal x(n), where we havereplaced the sampled time argument “n” with the continuous time argument“t”. The signal name “x” remains the same to indicate the sampled timeand continuous time signals represent the same signal, even though therewill generally be scaling differences and small time delays in the twosignal representations. This Nyquist sampling theory and practice ofsampled data systems is well-known to those skilled in the art.

A block diagram of the operating environment of an example RFID reader1000 using amplitude modulated reader-to-tag link communications 3000 isshown in FIG. 1. The reader 1000 is connected to one or more antennas2000, which radiate the modulated transmit signal 3000 to one or moretags 4000. Some types of RFID tags decode commands from the reader whichare encoded in the amplitude modulation of the reader's RF carriersignal. In some cases the modulation depth of the reader's RF carrier isspecified to be from 80% to 100%. Examples of this are the ISO 18000-63protocol and EPCGlobal C1G2 protocol, also informally known as “Gen2”.

As with some other digital signaling techniques, RFID information isrepresented in data-1 and data-0 intervals. Gen2 encodes commands to thetags using pulse interval encoding (PIE). In PIE, the information iscontained in the time interval between pulses. For Gen2, the protocoluses a short pulse interval for a data-0 and a longer pulse interval fora data-1. Still longer intervals are used for other special symbolsknown in the protocol as rtcal and trcal. The rtcal symbol is used totell the tag what time interval threshold to use for decoding data-0verses data-1. The trcal symbol is used to tell the tag what subcarrierfrequency to use when responding to the reader's commands. There is alsoa start-of-command symbol called delimiter which is simply a pulse offixed width approximately 12.5 microseconds long. Because of the passiveand semi-passive nature of most Gen2 tags, a “pulse” is defined as ashort, deep modulation of the RF carrier. In other words, a Gen2 pulseis a brief absence of the RF carrier. The duration of the short data-0interval is also sometimes referred to as the type A reference interval,or “tari”. Some figures in this disclosure contain time domain plotswhich are normalized by tari for clarity.

The previous paragraph refers to encoding in Gen2, or “RAIN”, as it isnow commonly referred to in the market. To improve the data rate beyondGen2 or RAIN, there are many modulations and encodings which can be usedto transmit information using amplitude modulation, such as M-ary AM.M-ary AM can be combined with position, interval, or angle modulation tocreate more complex symbols to convey even higher data rates.Information can be contained in the amplitude pulse levels; amplitudepulse position, timing, or intervals; amplitude phase reversals; or anycombination of the previously mentions parameters. Techniques to improvebandwidth efficiency are known to those skilled in the art. Thesetechniques include 4-ary AM to send 2 bits per symbols, 8-ary AM to send3 bits per symbol, and so forth which can be readily combined with pulseinterval or position encoding for even higher data rates. Anglemodulation may also be included to provide additional informationbearing parameters for RFID tags that may be capable of decoding anglemodulation. For passive RFID tags relying on amplitude modulation fordecoding there must be an AM component, even if additional informationis contained in the position, interval, frequency, phase, or otherparameters. This specification therefore universally refers to “AMtransmit signals” even though there could also be phase, frequency,position, interval, or any number of other parameters being modulated.The orthogonal offset technique disclosed herein is used to control themodulation depths of AM transmit signals.

Note that although the example embodiments described herein are focusedon the Gen2 protocol for ultra-high frequency (UHF) in the 900 MHzregion, the methods and apparatus described in the following can readilybe applied to high frequency (HF) RFID readers in the 13.56 MHz band, orto the 2.45 GHz microwave band, or other RFID readers in the HF, UHF, ormicrowave bands. The principal commercial application being addressedherein is efficient implementation of amplitude modulated reader-to-tagcommunications. Amplitude modulation is commonly used in RFID where thetags are very inexpensive and use envelope detection to decodecommunications from the reader. The modulated reader-to-tag signal 3000may also contain phase modulation, but many types of tags only use theenvelope for decoding.

FIG. 2 illustrates an architecture for RFID reader 1000 using a directconversion transmitter to generate the amplitude and phase modulated RFsignal. In this example the in-phase 1202 and quadrature-phase 1204baseband signals, u_(I)(n) and u_(Q)(n), respectively, are produceddigitally in software and/or hardware using the digital signal processor(DSP) 1100. The baseband signals 1202 and 1204 are inputs to thedigital-to-analog converters (DACs) 1302 and 1304 respectively, whichproduce baseband analog output signals. FIG. 2 shows an optional DCoffset circuit 1700, which may serve as the orthogonal offset generatorand in this example includes in-phase and quadrature-phase DC offsetsignals v_(I)(m) 1702 and v_(Q)(m) 1704, respectively, converted toanalog signals through DACs 1712 and 1714, respectively. Signals 1702and 1704 may also serve as transmitter nulling signals or can serve forboth nulling and orthogonal offset generation. A different samplingindex m is used since the sampling rate of 1702 and 1704 will typicallybe different from the main baseband signals 1202 and 1204. The analog DCoffset signal outputs from DACs 1712 and 1714 are inputs to the optionalDC offset summation circuits 1722 and 1724 which add the DC offsets tothe main transmit baseband signals from DACs 1302 and 1304,respectively. The DC offset signals and summation circuitry can have twopossible functions. One function may be to trim out nuisance DC offsetsand RF carrier feed through in the baseband analog circuitry andquadrature mixer. This function is conventional and known to thoseskilled in the art. An additional function according to some embodimentsof the disclosure herein may be to sum in an intentional DC offsetorthogonal to the PR-ASK signal modulation. This creates a newmodulation format this disclosure will denote as offset phase reversalamplitude shift keying (OPR-ASK). This will be described in great detailbelow.

The transmit baseband signals are passed through low pass filters (LPF)1322 and 1324 to produce the final baseband modulation signals z_(I)(t)1332 and z_(Q)(t) 1334, respectively. Note that the filters 1322 and1324 could precede the summing junctions 1722 and 1724 in alternativeembodiments since the DC offsets are typically constant signals with nomodulation. The signals 1332 and 1334 are baseband inputs to quadraturemodulator 1340, which also gets local oscillator input cos(ωt) from theRF source 1410, which provides the carrier. The quadrature modulatorideally creates the signalz _(ideal)(t)=z _(I)(t)cos(ωt)+z _(Q)(t)sin(ωt).

and ideally the inputs 1332 and 1334 to the quadrature modulator areperfect continuous time replicas of the digital signals 1202 and 1204,respectively, plus the optional DC offset signals v_(I)(m) 1702 andv_(Q)(m) 1704, respectively. Note that continuous time signals use timevariable t as the argument while sampled time signals use k, m, or n asthe argument. The continuous time representation of u_(I)(n) is denotedu_(I)(t), and the band limited sampling theory which ties the twotogether is known to those skilled in the art.

Returning to FIG. 2, the output signal from quadrature modulator 1340passes into the RF power amplifier (RFPA) 1350 which produces a highpower version applied to the TX-RX coupler 1420. Ideally the RFPA 1350has a linear transfer function from input to output. However, all poweramplifiers exhibit some nonlinearity. The nonlinearity can causeincreased spectral occupancy and adjacent channel interference problems.The nonlinearity is often characterized by two functions: amplitudedistortion and amplitude-to-phase distortion. Amplitude distortiontypically has a more significant effect on the spectral regrowthphenomenon and the degradation of spectral occupancy in the transmitsignal. Amplitude distortion is the effect wherein the gain of theamplifier depends on the input drive level. This is also sometimesreferred to as gain compression and gain expansion (both of which can beexhibited on the same device). Amplitude-to-phase distortion is asecondary problem, which also must be mitigated for high performancesystems. Amplitude-to-phase distortion is the phenomenon wherein thephase shift through the RFPA varies depending on the input drive level.A common technique to mitigate the problem of amplitude and phasedistortion is digital predistortion in the DSP 1100. This will bediscussed further in subsequent paragraphs.

Continuing with FIG. 2, the coupler 1420 sends substantially most or allof the high power transmit signal to the antennas, while any signalcoming into the reader from the antennas is mostly or all passed intothe receiver 1500. The receiver typically produces in-phase andquadrature-phase baseband receiver outputs which are inputs toanalog-to-digital converters (ADCs) 1602 and 1604. The sampled andquantized baseband receiver output is passed into the DSP 1100 forprocessing and decoding. The DSP 1100 interfaces with a client device toreport tag responses.

FIG. 3 shows a plot of a conventional PR-ASK modulated command signal inan RFID system. FIG. 3A shows an example PR-ASK baseband waveform 3110using a type A reference interval, or tari, of 12.5 microseconds and adata-1 length of 1.88×tari, or 23.5 microseconds. In FIG. 3 the timeaxis has been normalized to tari. The DSP generated PR-ASK signal willbe denoted s(n) or s(t), depending on whether it is being referenced inthe digital or analog domain, respectively. The signal 3110 is thein-phase part of the conventional PR-ASK signal in this exampleembodiment, and the quadrature-phase signal part 3120 is also shown inFIG. 3A. In this example the quadrature-phase part 3120 is exactly zero.Without loss of generality this disclosure will always normalize thesignal amplitude to one since the overall gain structure of the RFIDreader's transmitter is not important with respect to this new transmitsignal modulation method.

FIG. 3B shows the PR-ASK signal envelope 3130, which represents theenvelope of the RF signal transmission 3000 from the RFID reader 1000.Passive RFID tags 4000 decode the reader commands from the envelope ofthe RFID reader transmission. The PR-ASK signal envelope 3130 is fullymodulated since under ideal conditions with nuisance offsets fullycancelled the envelope goes all the way to zero which produces 100%modulation depth. Under realistic conditions it is impossible toperfectly cancel nuisance offsets and some residual offset will remainon the in-phase and quadrature-phase baseband circuitry and due to themixer 1340. These imperfections will cause the modulation depth to beless than 100%, although we will still refer to PR-ASK as being a fullymodulated signal since the intention of conventional PR-ASK is 100%amplitude modulation depth. Another common way to specify amplitudemodulation depth is as the ratio in decibels of full scale to minimumamplitude. Full scale in our examples is 1, which is the referenceamplitude our example signals are scaled to. In practice it is acceptedthat the modulation depth will should be near 30 dB or better, with 30dB representing 10^(−30/20)=0.0316 residual envelope, or about 97%amplitude modulation. In spite of these imperfectly un-cancelled offsetsthis specification will refer to these PR-ASK implementations as “fullymodulated”.

FIG. 3C shows the PR-ASK signal phase 3140 which is seen to jump betweentwo values, 0 degrees and 180 degrees, as the baseband signal 3110crosses back and forth through zero going positive and negative. FIG. 3Crepresents the PR-ASK phase under ideal conditions with nuisance offsetsfully cancelled. As discussed in the previous paragraph, in conventionalPR-ASK there is frequently residual un-cancelled offsets in the DACs,baseband circuitry, and mixer 1340 which affect the signal phasecharacteristics. The effect of offsets on the phase depends on themagnitude and polarity of the uncontrolled offsets on both the in-phaseand quadrature-phase circuitry. The phase may or may not have abrupt 180degree phase jumps, but it will nevertheless have extremely high slewrates in terms of degrees per second. Passive RFID tags typically do nothave local oscillators with which to coherently demodulate the RFIDtransmission, so phase modulation is transparent to the RFID tags.However, such high slew rates and/or abrupt phase jumps make polartransmitter implementations impractical.

In alternative embodiments of the conventional PR-ASK signal generationthe in-phase and quadrature-phase parts 1202 and 1204, respectively, maybe approximately equal to one another. In other words, the DSP 1100output to the transmit DACs 1302 and 1304 could be u_(I)(n)=√{squareroot over (2)}·s(n) and u_(Q)(n)=√{square root over (2)}·s(n), where the√{square root over (2)} factor is merely to maintain the normalizedamplitude as previously noted. This implementation can be advantageousto maximize the utilization of the transmit DAC circuitry.

Here we introduce a useful signal representation we will term the phaseplane, which is a representation of the baseband transmit signal orbaseband equivalent transmit signal as a parametric equation of thein-phase and quadrature-phase components as a function of time, eithercontinuous time t or sampled time n. For example,x=u _(I)(t) and y=u _(Q)(t),which produces plots as shown in FIG. 4. In the parametric equationsabove “x” refers to the in-phase, or real part, and “y” refers to thequadrature-phase, or imaginary part of the complex signal. Withoutconsidering the effects of baseband digital predistortion, the curvesdescribed by the parametric equations above are straight lines in thephase plane as shown by signal trajectories 3100 and 3200. Thetrajectory 3100 in FIG. 4 is the phase plane plot of the PR-ASK signalshown in FIG. 3A. The trajectory 3200 is the phase plane plot of thesignal when 1302 and 1304 areu _(I)(n)=√{square root over (2)}·s(n), andu _(Q)(n)=√{square root over (2)}·s(n),as briefly mentioned in the preceding paragraph. This is seen to resultin the trajectory on a θ=45 degree angle, but the length, or amplitude,is the same as 3100 and the signal trajectory still ideally passesthrough zero in the phase plane, which leads to the fully modulated 100%modulation depth and a phase signal with 180 degree jumpdiscontinuities. More generally, any angle θ can be used for thetrajectorys ^(θ)(n)=s(n)·e ^(jθ),where j=√{square root over (−1)}. This equation simply rotates thetrajectory angle in the phase plane but the envelope remains the same asin FIG. 3B and the phase signal is the same as in FIG. 3C except for aphase offset of θ.

As mentioned in the preceding paragraph, the phase plane trajectoriesare straight lines when signal predistortion is not considered. Whenbaseband digital predistortion is used the effects is that the phaseplane trajectory would generally not travel on a straight line segmentbut on some curve which is dependent on the characteristics of the RFPAdistortion. If offset nulling and trimming were included the trajectorywould be shifted slightly depending on the characteristics of thenuisance offsets in the baseband and mixer circuitry. If gain and phaseimbalance compensation is included this would further alter the basebandsignal trajectory. Note that offset nulling and gain/phase imbalanceequalization can be referred to as linear predistortion, and nonlinearpredistortion and linear predistortion can sometimes just be referred toas predistortion. These effects are not included in FIG. 4 for clarity.However, if nonlinear predistortion and/or linear predistortion wereincluded in the outputs to the transmit DACs, it does not depart fromthe disclosure herein. Indeed, the goal of digital predistortion is toinvert the effects of the offsets, gain and phase mismatches, andamplitude and phase distortion in the analog circuitry so that the finalhigh power transmitter output signal represents the ideal signalmodulation as closely as possible, and therefore the phase planetrajectory of the transmitter output baseband equivalent representationis very close to a straight line because the effects of the analogcircuit imperfections are ideally cancelled by the baseband digitalpredistortion.

In the conventional PR-ASK modulation scheme described above at anyangle θ, the PR-ASK signal trajectory passes through zero in the phaseplane creating a number of problems for the RFID reader system design.Many RF power amplifier (RFPA) architectures exhibit significantnonlinear behavior as the signal trajectory passes near the phase planeorigin. Furthermore, attempts to linearize the RFPA using basebanddigital predistortion inside the DSP can be complicated by estimatingthe amplitude and phase distortion characteristics at very low signallevels as the signal trajectory goes near the phase plane origin.Conventional designs use a transmitter offset nulling generator tominimize nuisance offsets in the baseband and mixer circuitry, whichcommonly can reduce the offsets to well under 5% of the full scalesignal. Frequently the transmitter offset nulling generator performswell enough for the fully modulated PR-ASK signal to have 99% modulationdepth, or a modulation depth of 40 dB (since 10^(−40/20)=0.01).

FIG. 5 shows a plot of one embodiment of a modulation format denoted asoffset phase reversal amplitude shift keying (OPR-ASK). The PR-ASKmodulated signal s(n) in an RFID system is scaled and offset from theorigin in the signal phase plane by adding a small orthogonal constantvalue whose magnitude is denoted as “B”, resulting ins _(B)(n)=s(n)·√{square root over (1−B ²)}+jB.The in-phase portion of s_(B)(n) is scaled by √{square root over (1−B²)}so that we maintain the unity normalization. FIG. 5A shows an exampleOPR-ASK baseband waveform with in-phase 3310 and quadrature-phase 3320signals. As in FIG. 3, the FIG. 5 time axis has been normalized to tari.FIG. 5B shows the envelope 3330 of the example OPR-ASK waveform in FIG.5A. FIG. 5C shows the signal phase 3340 for the example OPR-ASK waveformin FIG. 5A.

This example clearly illustrates two significant advantages of OPR-ASKover conventional PR-ASK. FIG. 5B shows that the OPR-ASK signal envelopenever goes close to zero. The Gen2 specification calls for a minimummodulation depth of 80%. The OPR-ASK modulation depth is given byM=1−B,which means that B≤0.2 to comply with the Gen2 specification. Unlikeconventional PR-ASK, the new OPR-ASK modulation allows the RFID designengineer choice in the modulation depth via the parameter 0<B≤0.2. Thesecond advantage of OPR-ASK is that the phase modulation is continuous,as illustrated in FIG. 5C. Unlike the phase signal of PR-ASK, which hasjump discontinuities as illustrated in FIG. 3C, the phase signal 3340varies smoothly and continuously between its two extreme values. Thetotal range of phase modulation is always less than 180 degrees. Forlarge values such as 0.1≤B≤0.2, the transmitter nulling offset generatorcan reduce nuisance offset to levels such that they have minimal impacton the phase and envelop signals of the OPR-ASK signal.

FIG. 6 shows the phase plane trajectory 3300 for the OPR-ASK examplesignal illustrated in FIG. 5A. It can be seen that the signal trajectory3300 is offset from the real axis by B. As with conventional PR-ASK, theOPR-ASK signal trajectory can be at any arbitrary angle θ. The equationbelow illustrates how to rotate the OPR-ASK signal s_(B)(n) to any angleθ,s _(B) ^(θ)(n)=s _(B)(n)·e ^(jθ)=(s(n)·e ^(jθ)·√{square root over (1−B²)})+j(B·e ^(jθ)).FIG. 6 shows the phase plane trajectory for the example OPR-ASK signalon a 45 degree angle 3400. Note that the signal trajectory remains aminimum distance of B away from the origin.

It is important for the offset B to be approximately orthogonal to thetrajectory of the PR-ASK signal modulation component in the phase plane.This ensures that the amplitudes of the signal symbol waveforms areapproximately equal which is a requirement of the Gen2 and ISO 18000-63specification. If the offset B is not substantially orthogonal to thesignal modulation, then signal envelope levels would alternate inamplitude and degrade the decoding margin for the tags, and may violatethe Gen2 specification even for moderately small offsets B.

There are significant advantages to the OPR-ASK modulation technique.Many high efficiency RFPA architectures, such as deep class-AB andclass-C amplifiers, exhibit significant amplitude and phase distortionfor signal levels near zero. Conventional PR-ASK modulation is 100%modulation depth and will always produce very low signal levels.Furthermore, if baseband digital predistortion is used then the newOPR-ASK modulation format makes it much easier to characterize the RFPAnonlinearity as compared to conventional PR-ASK. This is primarilybecause the amplitude and phase distortion does not need to be measurednear the origin in the phase plane. Finally, the spectral occupancy ofOPR-ASK modulation is typically less than conventional PR-ASK. To seethis, let the power spectral density of PR-ASK be denoted S(f). Then itis easy to see that the power spectral density of OPR-ASK isS _(B)(f)=(1−B ²)S(f)+B ²·δ(f),where δ(f) is the Dirac delta function. Thus, the introduction of theorthogonal offset moves a small fraction of the OPR-ASK signal power tof=0, in baseband equivalent representation, or to the RF carrierfrequency when considering the passband representation. Assuming themaximum value of B=0.2, the (1−B²) scaling factor on the modulated powerspectrum component S(f) above represents a 0.18 dB reduction. Therefore,the modulated power spectrum is slightly reduced in OPR-ASK as comparedto PR-ASK. The main technical and commercial advantages of OPR-ASK comefrom its reduced modulation depth and its continuous phase modulation.The reduced modulation depth reduces the nonlinear distortion producedby power efficient RFPAs and also makes it easier to implement nonlinearpredistortion, both of which lead to improved spectral occupancy byreducing spectrum regrowth in the transmitter system. The continuousphase modulation enables polar transmitter architectures for Gen2 RFIDwhich were not practical for PR-ASK signals due to the phasediscontinuities.

Using OPR-ASK, a system designer can, as an example use 80% to 95%modulation depth by selecting parameter 0.05<=B<=0.2. For purposes ofthis disclosure, this modulation depth is less than a “fully modulated”signal because the modulation depth has been intentionally andspecifically limited by a controlled orthogonal offset; the reducedmodulation depth is not due to impairments in the radio electronicsand/or nulling offset generator limitations. A good reader design willstill be able to cancel the nuisance offsets so that they are smallrelative to the controlled orthogonal offset. Fully modulated PR-ASKsignals ideally exhibit 100% modulation, but sometimes the modulationlevel may be as bad as 97% due entirely to poor offset cancellation.This is undesirable since the less-than-100% modulation depth is due tounintentional offsets whose magnitude and phase are unknown and mayadversely affect the tags' abilities to decode symbols due to imbalancebetween positive and negative waveforms. OPR-ASK modulation is reducedamplitude modulation depth by design using a controlled orthogonaloffset, which would usually be selected to be larger than residualnuisance circuitry offsets. This range of modulation depth keeps theorthogonal offset large relative to un-cancelled nuisance offsets.

FIG. 7 shows a block diagram of one embodiment of the disclosure on DSP1100. The RFID signal transmission sequence is encoded in the digitalsignal d(m) 1102. The digital signal d(m) may be a pulse code modulatedrepresentation or some other discrete representation of the RFID signaltransmission. Signal 1102 is an input to the PR-ASK signal generator1104, which produces a scaled PR-ASK signal 1106, √{square root over(1−B²)}·s(n). The sample index n at the output is different from thesample index m at the input because the sampling rates are generallydifferent. As discussed previously the signal 1106 is scaled so that theenvelope resulting after orthogonal offset addition of B will benormalized to unity. This is essentially for ease of analysis, and inpractice the signals will be scaled to optimize signal-to-quantizationnoise given the word width on the DSP and/or scaled to optimize othercriteria. Those skilled in the art understand the design criteria andtradeoffs involved in DSP systems.

The PR-ASK signal generator 1104 may be based on conventional Nyquistfiltering techniques or it may be based on optimization algorithms suchas those shown in the PCT Patent Application entitled, “WaveformSynthesis for RFID Transmitters,” Serial Number PCT/US2013/074897, filedon Dec. 13, 2013, incorporated herein by reference. Signal 1106 may bedirectly output as in-phase component 1202 to the DAC, while the offsetB 1190 is output on the orthogonal channel 1204. This is the mode ofoperation illustrated in FIG. 5A. FIG. 7 shows an optional rotation 1111consisting of four multiplications 1A, 1B, 1C, 1D, and two additions 1E,1F. The weights of the multiplications depend of the desired rotationangle θ and are generated from Euler's formulae ^(jθ)=cos θ+j sin θ, therefores _(B) ^(θ)(n)=s _(B)(n)·e ^(jθ)=(√{square root over (1−B²)}·s(n)+jB)(cos θ+j sin θ)s _(B) ^(θ)(n)=(√{square root over (1−B ²)}·s(n)cos θ−B sinθ)+j(√{square root over (1−B ²)}·s(n)sin θ+B cos θ)From which we see that multiplier 1A has weight cos θ, multiplier 1B hasweight sin θ, multiplier 1C has weight −sin θ, and multiplier 1D hasweight cos θ. The outputs of adders 1E and 1F are the real and imaginaryparts of s_(B) ^(θ)(n), 1108 and 1109, respectively. Note that if norotation is desired, θ=0, then the multiplier weights reduce to1A=1D=unity and 1B=1C=zero. This is the same as simply omitting thesecomponents for θ=0.

Continuing with FIG. 7, digital predistortion block 1120 may optionallybe included to apply predistortion to the signal s_(B) ^(θ)(n). NuisanceDC offset and carrier feed through trimming can optionally be done byadding small additional offset component, as shown by the nullingsubsystem 1130 in FIG. 7. Optional summation components 1132 and 1134together with the offset trim estimates 1136 serve as a transmitternulling offset generator to remove unwanted carrier feed-through.Alternatively, nuisance DC offset and carrier feed through trimming canbe done by using auxiliary DACs 1712 and 1714 as shown in FIG. 2, whichis not shown in FIG. 7. In either case, digital signals 1202 and 1204are passed out from the DSP 1100 to the transmit DACs 1302 and 1304. Itis also common to optionally perform gain and phase imbalanceequalization on the baseband signal. This is not shown in FIG. 7 but iswell-known to those skilled in the art. When nonlinear predistortion1120 is performed the gain/phase equalization must be done after thenonlinear predistortion 1120, although it may be placed before or after1130.

An alternate embodiment of the disclosure is to use the optional DACs1712 and 1714 of FIG. 2 to add in the orthogonal offset. In this casethe signals from FIG. 2 would beu _(I)(n)=√{square root over (1−B ²)}·s(n)cos θ  Signal 1202:u _(Q)(n)=√{square root over (1−B ²)}·s(n)sin θ  Signal 1204:v _(I)(m)=−B sin θ  Signal 1702:v _(Q)(m)=B cos θ  Signal 1704:Signals 1702 and 1704 may also include transmitter nulling offsetssummed in as desired. This embodiment has several disadvantagesincluding extra analog circuitry, less precision in the orthogonaloffset due to imperfections in the analog circuitry, and a morecumbersome digital predistortion implementation, if predistortion isused.

Note that any modulation angle θ in the phase plane is easilyaccommodated with the direct conversion architecture of FIG. 2. Thefundamental idea is that a specific, intentional (“controlled”) offsetis added orthogonally to the conventional PR-ASK signal so that thephase plane signal trajectory stays away from the origin. The offsetorthogonality ensures that the two extremes of the OPR-ASK signal haveapproximately the same amplitude. This OPR-ASK has the advantages overconventional PR-ASK previously discussed. Because the added offset isorthogonal to the PR-ASK component it is uncorrelated with the PR-ASKand it simply adds the carrier spectral component. Others skilled in theart may find alternative embodiments for injecting a carrier componentwhich is essentially orthogonal to the PR-ASK signal modulation. Forexample, it may be possible to sum in an orthogonal carrier offsetwithin the radio RF circuitry. Such alternative embodiments are stillwithin the spirit of this disclosure and should be considered coveredwithin this specification.

FIG. 8 shows a block diagram of DSP 1100 to produce amplitude 1203 andphase 1205 signals, a(n) and p(n) respectively, for an alternateembodiment of the disclosure applied to a polar transmitter RFID reader.Additional discussion of a polar transmitter can be found in theaforementioned Patent Application PCT/US2013/074897, and furtherdiscussion of polar transmitters can be found in the PCT PatentApplication PCT/US2014/068488 entitled, “Polar Transmitter for an RFIDReader,” filed in the U.S. Receiving Office on Dec. 4, 2014, claimingpriority from U.S. Provisional Patent Application 61/937,789, filed onFeb. 10, 2014, all incorporated herein by reference. In FIG. 8, the RFIDsignal transmission sequence encoding d(m) 1102 is input to the PR-ASKsignal generator 1104 to produce output 1106. Summer 1150 adds thequadrature phase offset jB 1192 to provide the orthogonal offset andcreate the new OPR-ASK modulation signal, which is then passed as inputto absolute value function 1160 and angle calculation function 1170.Note the output of summer 1150 is a complex signal which is representedin the block diagram as a wide arrow as in 1155. The functions 1160 and1170 produce the amplitude and phase signals, respectively. The DSP mayoptionally include a combined digital predistortion and offset trimmingblock 1140. The offset trimming provides a transmitter nulling offset.The final amplitude 1203 and phase 1205 signals are sent to the transmitDACs. A variation on the embodiment shown in FIG. 8 would be to move thedigital predistortion processing before the 1160 and 1170 components.This does not depart from the ideas conveyed herein.

A signal generation technique wherein only one polarity of a waveform isstored and a polarity generator is used cannot be used directly with theorthogonal offset incorporated into the stored waveforms. This isbecause the polarity generator, both for Cartesian based directconversion radios and polar modulation radios would incorrectly modifythe orthogonal offset. Thus, the example embodiments illustrated here inFIG. 7 and FIG. 8 implement nonlinear and linear predistortion asseparate functional blocks after the orthogonal offset has been added.The nonlinear predistortion cannot happen before the orthogonal offsetis added since the offset affects the drive level into the RFPA. Linearpredistortion is typically used to mitigate offsets, gain and phaseimbalance in the baseband and mixer, and carrier feed through in themixer. These linear impairments occur in the baseband and mixer beforethe signal reaches the RFPA where the dominant nonlinear distortionoccurs. The linear predistortion, to mitigate the baseband and mixerlinear impairments, must therefore occur after the nonlinearpredistortion. Given this ordering of operations:

(1) adding a controlled orthogonal offset,

(2) nonlinear predistortion, and

(3) linear predistortion (offset and gain/phase imbalance equalization),the predistortion operations cannot be combined in the stored waveformsynthesis using only a single reference waveform for each symbol. ForOPR-ASK, both the reference waveform and the phase reversed waveformmust be stored for each symbol. The following figures and text discloseconcepts of how to implement stored waveform transmit synthesistechniques which are also capable of integrating the nonlinear andlinear predistortion into the stored waveforms, even when using theOPR-ASK modulation technique.

FIG. 9 is a block diagram of a system for DSP 1100 wherein two alternatephases of the symbol waveforms are stored in a storage medium, memory5000 associated with DSP 1100. The memory could be internally integratedinto DSP 1100, or external volatile or nonvolatile memory that DSP 1100has access to. There are seven waveforms shown in FIG. 9:

w₀₊(n), 5110, the data-0 waveform associated with the positive OPR-ASKdata-0 symbol;

w₁₊(n), 5120, the data-1 waveform associated with the positive OPR-ASKdata-1 symbol;

w_(del+)(n), 5130, the delimiter waveform associated with the positiveOPR-ASK delimiter symbol;

w_(rtcal+)(n), 5140, the rtcal waveform associated with the positiveOPR-ASK rtcal symbol;

w_(trcal+)(n), 5150, the trcal waveform associated with the positiveOPR-ASK trcal symbol;

w_(stop+)(n), 5160, the stop waveform associated with the positiveOPR-ASK stop symbol; and

w_(cw+)(n), 5170, the continuous wave (CW) waveform associated with thepositive OPR-ASK CW symbol.

Note that there is no “stop” data symbol defined in the Gen2specification, but it may be implied in that a final rising edge of theRF envelope is needed to define the final data symbol within areader-to-tag command. It must also be pointed out that the terms“positive” and “negative” with regard to OPR-ASK modulation must begeneralized from their normal meaning. Positive and negative areproperties of real numbers. The PR-ASK signal 1106 is real valued where“positive” and “negative” waveforms take their usual meanings. SinceOPR-ASK is a complex valued modulation technique, we must generalizethese meanings. When this disclosure refers to a “positive” OPR-ASKwaveform it is meant the portion of the waveform resulting from thepositive scaled PR-ASK signal 1106, possibly including zero as well.When this disclosure refers to a “negative” OPR-ASK waveform it is meantthe portion of the waveform resulting from the negative scaled PR-ASKsignal 1106, possibly including zero as well. This disclosure will alsouse the term “reference” for the positive waveform and “reversed” or“reversed phase” to refer to the negative version.

As shown in FIG. 9, the DSP memory 5000 also contains waveformsassociated with phase reversed OPR-ASK waveforms as follows:

w⁰⁻(n), 5210, the data-0 waveform associated with the reversed OPR-ASKdata-0 symbol;

w¹⁻(n), 5220, the data-1 waveform associated with the reversed OPR-ASKdata-1 symbol;

w_(del−)(n), 5230, the delimiter waveform associated with the reversedOPR-ASK delimiter symbol;

w_(rtcal−)(n), 5240, the rtcal waveform associated with the reversedOPR-ASK rtcal symbol;

w_(trcal−)(n), 5250, the trcal waveform associated with the reversedOPR-ASK trcal symbol;

w_(stop−)(fn), 5260, the stop waveform associated with the reversedOPR-ASK stop symbol;

w_(cw−)(n), 5270, the continuous wave (CW) waveform associated with thereversed OPR-ASK CW symbol.

More generally, there may be multiple versions of the waveformsdepending on the variety of link configurations supported for the Gen2air interface. There may be multiple versions of the delimiter anddata-0 symbols depending on the frame sync implementation and linkconfigurations. These variations do not depart from the ideas disclosedherein.

As shown in FIG. 9, the RFID transmit information is encoded in a symbolsequence d(k) 6010, which the index k is used here to indicate adiscrete sequence. The sampling rate of this sequence is typicallynon-uniform because of the different lengths of the RFID commands. TheDSP 1100 has one or more mux logic apparatus which use the symbolswithin the sequence 6010 to select from a plurality of waveforms to readand output from memory 5000. FIG. 9 shows two multiplexers, 6110 and6120. In the example implementation of FIG. 9 multiplexer (mux) 6110selects which of the positive reference phase waveforms to output basedon the current symbol d(k), while mux 6120 selects which of the reversedreference phase waveforms to output based on the current symbol d(k). Inthis embodiment the DSP has a phase selection switch apparatus 6300which performs an analogous function as the previously discussedpolarity generator. The phase select switch 6300 alternates between themux 6110 output and mux 6120 output every new symbol, except if the newsymbol is a delimiter or a CW symbol, in which case the phase selectswitch does not alternate. The mux implementation has a record of howlong each waveform symbol is, and after the complete waveform has beentransferred from memory a symbol clock is provided so as to make thesystem move on to the next symbol in the sequence 6010. FIG. 9 showseach mux 6110 and 6120 providing a symbol clock output, 6210 and 6220,respectively. There are a significant number of ways to implement thiswhether it is done in software, in hardware logic gates, or in someprogrammable gate array. FIG. 9 shows a block diagram of onepossibility.

Continuing with FIG. 9, phase select switch 6300 routes the correctphase waveform to the demultiplexer (demux) 6400. Note that wide arrowssuch as 6320 in FIG. 9 represent complex valued signals, whether inCartesian or polar representation. The demux 6400 separates the complexsignal representation into the component signals to be output to theDACs. The component signals f_(A)(n) 6502 and f_(B)(n) 6504 may beCartesian representation signals u_(I)(n) 1202 and u_(Q)(n) 1204 for usein a direct conversion radio such as FIG. 2, or the component signalsf_(A)(n) 6502 and f_(B)(n) 6504 may be polar representation signals a(n)1203 and p(n) 1205 for use in a polar modulation transmitterarchitecture. Therefore, while FIG. 9 refers to DSP 1100 as in thedirect conversion architecture, the system illustrated in FIG. 9 isequally applicable to polar transmitter architectures.

FIG. 10 shows plots of example waveforms which would be stored in memory5000 for data-0 and data-1 signals for a direct conversion Cartesianrepresentation. The time axis in each of the plots is normalized totari. In FIG. 10A the positive data-0 waveform is shown for a directconversion Cartesian representation, where signal 5112 is the in-phasepart and signal 5114 is the quadrature-phase part. In FIG. 10B the phasereversed data-0 waveform is shown for a direct conversion Cartesianrepresentation, where signal 5212 is the in-phase part and signal 5214is the quadrature-phase part. In FIG. 10C the positive data-1 waveformis shown for a direct conversion Cartesian representation, where signal5122 is the in-phase part and signal 5124 is the quadrature-phase part.In FIG. 10D the phase reversed data-1 waveform is shown for a directconversion Cartesian representation, where signal 5222 is the in-phasepart and signal 5224 is the quadrature-phase part. These signals areexamples for use with the direct conversion radio as in FIG. 2. Thesignals are modulated along a 45 degree axis with a 135 degreeorthogonal offset, such as in signal trajectory 3400 shown in FIG. 6.Other orientations of the modulation are equally possible. Other typesof waveforms such as rtcal, delimiter, etc., may be similarlyconstructed. The orthogonal offset used in FIG. 10 was chosen togenerate 80% modulation depth, though any modulation depth from 80% to100% is included within these concepts.

FIG. 11 shows plots of example waveforms which would be stored in memory5000 for data-0 and data-1 as would be used for a polar modulation radioarchitecture. The time axis in each of the plots is normalized to tari.In FIG. 11A the positive data-0 waveform is shown, where signal 5113 isthe amplitude part and signal 5115 is the phase part. In FIG. 11B thereversed phase data-0 waveform is shown, where signal 5213 is theamplitude part and signal 5215 is the phase part. In FIG. 11C thepositive data-1 waveform is shown, where signal 5123 is the amplitudepart and signal 5125 is the phase part. In FIG. 11D the phase reverseddata-1 waveform is shown, where signal 5223 is the amplitude part andsignal 5225 is the phase part. The signals are modulated along a 90degree axis with a 0 degree orthogonal offset. This orientation waschosen for ease of plotting since the phase signal is symmetric aboutzero in FIG. 11. In the plots of FIG. 11, the amplitude scale is on theleft half of the plots, while the phase scale (in degrees) is on theright side of the plots. Other types of waveforms such as rtcal,delimiter, etc., may be similarly constructed. The orthogonal offsetused in FIG. 11 was chosen to generate 80% modulation depth, though anymodulation depth from 80% to 100% is included within these concepts.

FIG. 12 is a block diagram of an alternative system for DSP 1100 whereina single mux 6130 is used to select the correct symbol waveformincluding which phase to read from memory. In FIG. 12, the transmitterencoding sequence d(k) 6020 may have phase information embedded in it sothat the single mux 6130 has enough information to select the correctwaveform. There are many possible embodiments of this concept. Threeexample embodiments are given here. In the first example, a unique codecan be assigned to each waveform, such as

-   -   0=data-0 positive reference phase    -   1=data-0 reversed reference phase    -   2=data-1 positive reference phase    -   3=data-1 reversed reference phase    -   4=delimiter positive reference phase    -   5=delimiter reversed reference phase    -   6=rtcal positive reference phase    -   7=rtcal reversed reference phase    -   8=trcal positive reference phase    -   9=trcal reversed reference phase    -   10=stop positive reference phase    -   11=stop reversed reference phase    -   12=CW positive reference phase    -   13=CW reversed reference phase.        In this example, the transmitter encoding sequence d(k) 6020 in        FIG. 12 would take on values between 0 and 13 to encode the RFID        reader system transmissions. With this type of explicit        enumeration any ordering is possible, and the above is just for        illustration. More or less waveforms and symbols can be used        without departing from the fundamental ideas disclosed herein.

In a second example encoding, the waveform type can be enumerated suchas

-   -   0=data-0    -   1=data-1    -   2=delimiter    -   3=rtcal    -   4=trcal    -   5=stop    -   6=CW.        Then an additional field, such as an additional bit location in        a binary encoded system, is used to encode the phase of the        symbol. Note the manner in which the first example above was        enumerated also behaves similar to this second example, since        the least significant bit in that binary encoding would        explicitly represent the phase. Other orderings of the symbols        in the first example would not have a “phase bit field” as        embodied in the second example.

In a third example encoding of the transmitter encoding sequence d(k)6020 in FIG. 12, the phase could be represented differentially, such asby a “reverse phase bit field”. The sequence encoder would set thisphase reverse field when the new waveform symbol needed to be phasereversed, and the phase reverse field would be clear when the phase didnot need to be reversed. The mux 6130 would have an internal phase statein this embodiment which could be preset to either positive or reversedphase as desired.

Continuing with FIG. 12, note that wide arrows such as 6330 in FIG. 12represent complex valued signals, whether in Cartesian or polarrepresentation. The demux 6400 separates the complex signalrepresentation into the component signals to be output to the DACs Notethat the demux 6400 may not be explicitly implemented, but could simplybe a function accomplished inherently through the DSP to DAC datatransfer mechanism. The component signals f_(A)(n) 6502 and f_(B)(n)6504 may be Cartesian representation signals u_(I)(n) 1202 and u_(Q)(n)1204 for use in a direct conversion radio such as FIG. 2, or thecomponent signals f_(A)(n) 6502 and f_(B)(n) 6504 may be polarrepresentation signals a(n) 1203 and p(n) 1205 for use in a polarmodulation transmitter architecture. Therefore, while FIG. 12 refers toDSP 1100 in the direct conversion architecture, the system illustratedin FIG. 12 is equally applicable to polar modulation transmitters.

FIG. 9, FIG. 10, FIG. 11, and FIG. 12 show how to implement stored tablebased transmit signal synthesis using the orthogonal offset techniquewhile still capable of incorporating nonlinear and linear predistortioninto the stored waveforms. Having the nonlinear and linear predistortionincorporated into the stored waveforms is very computationallyefficient. For clarity, the waveform plots of FIG. 10 and FIG. 11 do notinclude predistortion, but it should be apparent to those skilled in theart that it is possible and desirable to include the predistortion inthe stored waveforms so that the predistortion operation does not haveto be implemented separately.

Other embodiments are possible using the ideas presented here. Forexample, a sequence encoding could be used in FIG. 12 with no phaseinformation, but the mux 6130 could implement all phase control bymaintaining a phase state and implementing the logic to invert the phasestate on all new symbols except delimiter and CW. This is similar to thepolarity generation control in aforementioned patent applicationPCT/US2013/074897. Any such implementation is still within the spirit ofthis disclosure.

Hybrid waveform table based synthesis designs are possible which reducewaveform memory 5000 storage requirements by using particular modulationangle trajectories together with simple adjustments to the mux logic.For example, polar modulation along a θ=90 degree trajectory yieldidentical waveform envelopes but with phase symmetric about zero for thepositive and negative waveforms. Likewise, Cartesian modulation alongθ=0, 90, 180, or 270 degree trajectories yield symmetries in thein-phase and quadrature-phase components which may be exploited toreduce storage memory. One skilled in the art could suitably modify thephase reversal logic in the mux given the information disclosed herein.It should be noted however that these signal symmetries are typicallyeliminated when nonlinear baseband predistortion is included in thewaveform synthesis.

FIG. 13 shows a flowchart 7000 illustrating the process of creatingOPR-ASK waveforms. The flowchart starts at 7010. At block 7020 thePR-ASK waveform is generated. This can be done using conventionalNyquist filtering such as using pulse code modulation filtered by araised cosine, or the PR-ASK waveform can be generated using anoptimization algorithm such as quadratic programming. At block 7030 thePR-ASK signal is scaled as needed for the DSP numerical word size or thewidth of the bus interface to the transmit DAC. The orthogonal offset isalso added at block 7030. The result at the end of block 7030 is theOPR-ASK waveform. At block 7040 the OPR-ASK waveform is rotated as maybe desired to optimize transmit DAC word size or for other reasons. Atblock 7050 the OPR-ASK signal may optionally be converted to polarcoordinates for use in a polar transmitter. At block 7060 the processmay optionally add nonlinear and/or linear predistortion to compensatefor nonlinearities, offsets, and gain/phase imbalance in the analogsection of the transmitter. At block 7090 the process is finished.

FIG. 14 shows a flowchart 8000 illustrating the process of synthesizingOPR-ASK signals from stored waveforms. This example embodiment followsan implementation as in FIG. 12, where a single mux control blockmaintains the current polarity state information and determines whichversion, positive or negative, of a waveform should be sent to the DACs.The flowchart starts at 8010. At block 8012 the process checks if a newRFID transmitter command is ready to be sent in the encoded symbolsequence d(k). If no new command is ready, the process continues toblock 8014 where the mux selects the positive or negative CW waveformdepending on the current polarity state. From there the processcontinues to block 8016 where the waveform length counter is initializedto the length of the CW waveform. From here, the process continues toblock 8042 where it rejoins the main flow of the algorithm. Returning toblock 8012, if there is a new RFID command ready to send, the processcontinues to block 8020 where the next symbol is read from the symbolsequence d(k). At block 8022 the process checks if the new symbol is aCW symbol, and if so continues onto block 8032. However, if the newsymbol is not a CW symbol, the process continues onto block 8024 whereit check if the new symbol is a delimiter. If the new symbol is adelimiter the process moves to block 8032, while if it is not adelimiter the process continues to block 8026 where the current polaritystate is inverted. At block 8032 the waveform is selected based on thenew symbol d(k) and the current polarity state. In example softwareimplementations the waveform is selected by passing a pointer to thebeginning of the waveform in memory. At block 8034 the mux logicinitializes the waveform counter based on the length of the newlyselected waveform. At block 8042 samples are read from waveform memoryand at block 8046 the samples are written to the transmitter DACs. Atblock 8050 the sample counter is decremented and at block 8052 thecounter is checked to determine if there are more samples to send. Ifthere are more samples to send the process continues back to block 8042.On the other hand, if no more samples are in the current waveform, thenthe process continues to block 8060 where the symbol sequence d(k) ischecked for more symbols in the command. If there are more symbols inthe command, the process loops back to block 8020 to process the nextwaveform. However, if there are no more symbols in the command theprocess loops back to the start where it checks for a new command at8012, thereby starting the process over again.

FIG. 15 shows plots of example amplitude and phase signals forconventional 4-ary pulse amplitude modulated transmit signal in an RFIDsystem. In FIG. 15 the time axis has been normalized to the symbol rate.FIG. 15A shows the pulse code sequence 3510 taking random values of {−3,−1, +1, +3}, thereby encoding 2 bits per pulse. The pulse amplitudesequence 3510 is filtered to produce the AM transmit signal, whoseamplitude is shown in FIG. 3A as 3520. The 10-symbol example sequence3510 illustrated in FIG. 15A consists of −1, +3, +3, −3, +1, −3, +3, −3,−3, +1. As seen in FIG. 15A, the tag can decode the presence or absenceof a polarity reversal by the presence of a deep modulation troughbetween the symbols.

The amplitude signal is the baseband representation of the envelope ofthe RF signal transmission 3000 from the RFID reader 1000. Passive RFIDtags 4000 decode the reader commands from the envelope of the RFIDreader transmission. The AM transmit signal envelope 3520 is fullymodulated since under ideal conditions with nuisance offsets fullycancelled the envelope goes all the way to zero which produces 100%modulation depth. Under realistic conditions it is impossible toperfectly cancel nuisance offsets and some residual offset will remainon the in-phase and quadrature-phase baseband circuitry and due to themixer 1340. These imperfections will cause the modulation depth to beless than 100%, although we will still refer to AM transmit as being afully modulated signal since the intention of conventional AM transmitis 100% amplitude modulation depth. Another common way to specifyamplitude modulation depth is as the ratio in decibels of full scale tominimum amplitude. Full scale in our examples is 1, which is thereference amplitude our example signals are scaled to. In practice it isaccepted that the modulation depth should be near 30 dB or better, with30 dB representing 10^(−30/20)=0.0316 residual envelope, or about 97%amplitude modulation. Despite these imperfectly un-cancelled offsetsthis specification will refer to these AM transmit implementations as“fully modulated”.

FIG. 15B shows the phase of the signal which takes on values of 180degrees such as 3530 or 0 degrees as in 3540. The absolute phase isarbitrary, but this type of 180-degree phase discontinuity is common indeeply modulated AM RFID systems. Phase signals 3530 and 3540 representthe AM transmit phase under ideal conditions with nuisance offsets fullycancelled. As discussed in the previous paragraph, in conventional AMtransmit there is frequently residual un-cancelled offsets in the DACs,baseband circuitry, and mixer 1340 which affect the signal phasecharacteristics. The effect of offsets on the phase depends on themagnitude and polarity of the uncontrolled offsets on both the in-phaseand quadrature-phase circuitry. The phase may or may not have abrupt180-degree phase jumps, but it will nevertheless have extremely highslew rates in terms of degrees per second. Passive RFID tags typicallydo not have local oscillators with which to coherently demodulate theRFID transmission, so phase modulation is transparent to the RFID tags.However, such high slew rates and/or abrupt phase jumps make polartransmitter implementations impractical.

FIG. 16 shows plots of example amplitude and phase signals for OAM usingthe same 4-ary pulse amplitude modulated transmit signal as was used inFIG. 15. In the simplest implementation of OAM, an orthogonal offset isadded to the signal constellation, e.g., the OAM constellation is{−3,−1,1,3}+jB={−3+jB,−1+jB,1+jB,3+jB},or expressed mathematically asC _(m) =A _(m) +jB,where C_(m) is the OAM constellation point corresponding to AMconstellation point A_(m), j=√{square root over (−1)}, and B is theorthogonal offset. This offset raises the average power of this OAMconstellation. Typically, a small orthogonal offset is needed such as0.2 or less. For an offset of 0.2 the power increase is less than 0.04dB. The power increase can be normalized out several ways. Oneembodiment is normalizing each point of the constellation to itsamplitude, for example,C _(m)=sign(A _(m))·√{square root over (A _(m) ² −B ²)}+jB,This implementation normalizes each point to the same amplitude, butresults in a small distortion in Euclidean distance between points; lessthan 0.09 dB distortion for an offset bias of 0.2. Another preferredembodiment is

${C_{m} = {\left( {A_{m} + {jB}} \right) \cdot \frac{P_{AM}}{P_{OAM}}}},$where P_(AM) is the average AM constellation power and P_(OAM) is theaverage OAM constellation power. In this case, the Euclidean distance isuniform between OAM constellation points is uniform but reduced by asmall amount. Note that the power increase or Euclidean distancedistortion will increase with increasing orthogonal offset magnitude B.In practice, only a small offset magnitude is needed to overcome thesevere cutoff distortion common in high efficiency RF power stages.Another note is that the phase is arbitrary. This specification has useda real AM constellation {−3, −1, +1, +3} with an imaginary offset jB.However, any angle will work, so long as the offset is substantiallyorthogonal to the constellation.

Continuing with FIG. 16A, 3610 is the pulse code sequence, while 3620 isthe amplitude of the filtered OAM transmit signal. Note the reducedmodulation depth 3625. The injection or biasing of the AM signal withthe orthogonal offset to create the OAM signal reduces the modulationdepth. FIG. 16B shows the phase of the filtered OAM signal, 3630. ForOAM, the phase signal is not only continuous, but continuouslydifferentiable. This is in sharp contrast to the discontinuous phasesignals 3530 and 3540 for AM.

Unlike conventional AM transmitter, the OAM modulation described hereinallows the RFID design engineer choice in the modulation depth and phasebandwidth via the offset parameter B. There are significant advantagesto the OAM modulation technique. Many high efficiency RFPAarchitectures, such as deep class-AB, class-C, class-E, or switch modeamplifiers, exhibit significant amplitude and phase distortion forsignal levels near zero. Conventional AM transmit modulation is 100%modulation depth and will always produce very low signal levels.Furthermore, if baseband digital predistortion is used then the new OAMmodulation format makes it much easier to characterize the RFPAnonlinearity as compared to conventional AM transmit. This is primarilybecause the amplitude and phase distortion does not need to be measurednear the origin in the phase plane. Finally, the spectral occupancy ofOAM modulation is typically less than conventional AM transmit. To seethis, let the power spectral density of AM transmit be denoted S(f).Then it is easy to see that the power spectral density of OAM isS _(B)(f)=(1−B ²)·S(f)+B ²·δ(f),where δ(f) is the Dirac delta function. Thus, the introduction of theorthogonal offset moves a small fraction of the OAM signal power to f=0,in baseband equivalent representation, or to the RF carrier frequencywhen considering the passband representation. Assuming the maximum valueof B=0.2, the (1−B²) scaling factor on the modulated power spectrumcomponent S(f) above represents a 0.18 dB reduction. Therefore, themodulated power spectrum is slightly reduced in OAM as compared to AMtransmit. The main technical and commercial advantages of OAM come fromits reduced modulation depth and its continuous phase modulation. Thereduced modulation depth reduces the nonlinear distortion produced bypower efficient RFPAs and makes it easier to implement nonlinearpredistortion, both of which lead to improved spectral occupancy byreducing spectrum regrowth in the transmitter system. The continuousphase modulation enables polar transmitter architectures for Gen2 RFIDwhich were not practical for AM transmit signals due to the phasediscontinuities.

Using OAM, a system designer can, as an example use 80% to 95%modulation depth by selecting parameter 0.05<=B<=0.2. For purposes ofthis disclosure, this modulation depth is less than a “fully modulated”signal because the modulation depth has been intentionally andspecifically limited by a controlled orthogonal offset; the reducedmodulation depth is not due to impairments in the radio electronicsand/or nulling offset generator limitations. A good reader design willstill be able to cancel the nuisance offsets so that they are smallrelative to the controlled orthogonal offset. Fully modulated AMtransmit signals ideally exhibit 100% modulation, but sometimes themodulation level may be as bad as 97% due entirely to poor offsetcancellation. This is undesirable since the less-than-100% modulationdepth is due to unintentional offsets whose magnitude and phase areunknown and may adversely affect the tags' abilities to decode symbolsdue to imbalance between positive and negative waveforms. OAM modulationis reduced amplitude modulation depth by design using a controlledorthogonal offset, which would usually be selected to be larger thanresidual nuisance circuitry offsets. This range of modulation depthkeeps the orthogonal offset large relative to un-cancelled nuisanceoffsets.

FIG. 17 shows a block diagram of one embodiment of the disclosure on DSP1100. The RFID signal transmission sequence is encoded in the digitalsignal d(m) 1102. Signal 1102 is an input to the AM transmit signalgenerator 1105. AM signal generator 1105 may incorporate one of thepreferred embodiment power normalization discussed above. AM signalgenerator 1105 produces signal 1107, s(n). The sample index n at theoutput is different from the sample index m at the input because thesampling rates are generally different. The AM transmit signal generator1105 may be based on conventional Nyquist filtering techniques or it maybe based on optimization algorithms such as those shown in the PCTPatent Application entitled, “Waveform Synthesis for RFID Transmitters,”Serial Number PCT/US2013/074897. Recall again that 1105 is referred toas “AM signal generator” even though there could also be phase,frequency, position, interval, or any number of other parameters beingmodulated. The orthogonal offset technique disclosed herein is useful tocontrol the modulation depths of AM transmit signals.

Signal 1107 may be directly output as in-phase component 1206 to theDAC, while the offset B 1190 is output on the orthogonal channel 1207.This is the mode of operation illustrated in FIG. 17A. FIG. 17 shows anoptional rotation 1112 consisting of four multiplications 1A, 1B, 1C,1D, and two additions 1E, 1F. The weights of the multiplications dependof the desired rotation angle θ and are generated from Euler's formulae ^(jθ)=cos θ+j sin θ,therefores _(B) ^(θ)(n)=s _(B)(n)·e ^(jθ)=(s(n)+jB)(cos θ+j sin θ)s _(B) ^(θ)(n)=(√{square root over (1−B ²)}·s(n)cos θ−B sinθ)+j(√{square root over (1−B ²)}·s(n)sin θ+B cos θ)From which we see that multiplier 1A has weight cos θ, multiplier 1B hasweight sin θ, multiplier 1C has weight −sin θ, and multiplier 1D hasweight cos θ. The outputs of adders 1E and 1F are the real and imaginaryparts of s(n), 1110 and 1112, respectively. Note that if no rotation isdesired, θ=0, then the multiplier weights reduce to 1A=1D=unity and1B=1C=zero. This is the same as simply omitting these components forθ=0.

Continuing with FIG. 17, digital predistortion block 1120 may optionallybe included to apply predistortion to the signal s_(B) ^(θ)(n). NuisanceDC offset and carrier feed through trimming can optionally be done byadding small additional offset component, as shown by the nullingsubsystem 1130 in FIG. 17. Optional summation components 1132 and 1134together with the offset trim estimates 1136 serve as a transmitternulling offset generator to remove unwanted carrier feed-through.Alternatively, nuisance DC offset and carrier feed through trimming canbe done by using auxiliary DACs 1712 and 1714 as shown in FIG. 2, whichis not shown in FIG. 17. In either case, digital signals 1206 and 1207are passed out from the DSP 1100 to the transmit DACs 1302 and 1304. Itis also common to optionally perform gain and phase imbalanceequalization on the baseband signal. This is not shown in FIG. 17 but iswell-known to those skilled in the art. When nonlinear predistortion1120 is performed the gain/phase equalization must be done after thenonlinear predistortion 1120, although it may be placed before or after1130.

An alternate embodiment of the disclosure is to use the optional DACs1712 and 1714 of FIG. 2 to add in the orthogonal offset. In this casethe signals from FIG. 2 would beu _(I)(n)=√{square root over (1−B ²)}·s(n)cos θ  Signal 1206:u _(Q)(n)=√{square root over (1−B ²)}·s(n)sin θ  Signal 1207:v _(I)(m)=−B sin θ  Signal 1702:v _(Q)(m)=B cos θ  Signal 1704:Signals 1702 and 1704 may also include transmitter nulling offsetssummed in as desired. This embodiment has several disadvantagesincluding extra analog circuitry, less precision in the orthogonaloffset due to imperfections in the analog circuitry, and a morecumbersome digital predistortion implementation, if predistortion isused. Note that any modulation angle θ in the phase plane is easilyaccommodated with the direct conversion architecture of FIG. 2. Thefundamental idea is that a specific, intentional (“controlled”) offsetis added orthogonally to the conventional AM transmit signal so that thephase plane signal trajectory stays away from the origin. The offsetorthogonality ensures that the two extremes of the OAM signal haveapproximately the same amplitude. This OAM has the advantages overconventional AM transmit previously discussed. Because the added offsetis orthogonal to the AM transmit component it is uncorrelated with theAM transmit and it simply adds the carrier spectral component. Othersskilled in the art may find alternative embodiments for injecting acarrier component which is essentially orthogonal to the AM transmitsignal modulation. For example, it may be possible to sum in anorthogonal carrier offset within the radio RF circuitry. Suchalternative embodiments are still within the spirit of this disclosureand should be considered covered within this specification.

FIG. 18 shows a polar embodiment of the OAM transmitter 1100. After theorthogonal offset bias 1192 is summed with the AM transmit signal 1107,the OAM transmit signal 1156 is converted to polar coordinates viaseparate amplitude function 1160 and phase function 1170, or through asingle Cartesian-to-polar conversion such as a CORDIC processor, knownto those skilled in the art. Optional function 1140 does predistortionand offset trimming to compensate for analog transmitter hardwareimperfections. The transmitter output is the amplitude signal a(n) 1208and phase signal p(n) 1209.

FIG. 19 shows a flowchart 7001 illustrating the process of creating OAMwaveforms. The flowchart starts at 7011. At block 7021 the AM transmitwaveform is generated. This can be done using conventional Nyquistfiltering such as using pulse code modulation filtered by a raisedcosine, or the AM transmit waveform can be generated using anoptimization algorithm such as quadratic programming. At block 7031 theAM transmit signal is scaled as needed for the DSP numerical word sizeor the width of the bus interface to the transmit DAC. The orthogonaloffset is also added at block 7031. The result at the end of block 7031is the OAM waveform. At block 7041 the OAM waveform is rotated as may bedesired to optimize transmit DAC word size or for other reasons. Atblock 7051 the OAM signal may optionally be converted to polarcoordinates for use in a polar transmitter. At block 7061 the processmay optionally add nonlinear and/or linear predistortion to compensatefor nonlinearities, offsets, and gain/phase imbalance in the analogsection of the transmitter. At block 7091 the process is finished.

The example devices, methods, apparatus, and embodiments disclosed inthis specification are not the only possible implementations. Anyalternative implementation devised with the intention of shifting thetrajectory of the PR-ASK signal away from the origin through theaddition of a controlled offset is the same idea as disclosed herein. Analternative may be to use an offset which is not ideally orthogonal tothe PR-ASK trajectory, but is intentionally skewed by a small amountsuch that the envelope still passes the protocol conformancerequirements. Such an implementation of intentionally skewing the offsetso that it is not ideally orthogonal but just mostly orthogonal, asmeasured by the percentage projection onto the PR-ASK trajectory versesthe axis 90 degrees to the trajectory, would still be within the spiritand scope of the idea disclosed here. Yet another alternative embodimentwould be to drive a transmit filter, such as a raised cosine filter orsome other Nyquist filter well known to those skilled in the art, usinga complex pulse code modulated signal representing the PR-ASK signalwith an intentional offset. In other words, a designer could use asequence d(m) as the encoded PR-ASK signal and algebraically add theoffset to produce the sequence d(m)+jB, which can then be passed as theinput to the transmit filter. The implementation could optionally rotatethe input or the output of the transmit filter to put the trajectory onany desired angle. Because the transmit filter, the optional rotation,and the algebraic summation of the offset are all linear operations, theorder of these operations is not important to produce the same endresult. In any such implementation the sequence d(m) can be viewed asthe PR-ASK generator and the summation with the offset can be viewed asthe offset generator with respect to what is claimed with thisspecification.

The example devices and methods in this disclosure can achieve FCC andETSI regulatory compliance with low computational complexity. In someembodiments, a general-purpose processor such as a DSP, microcontrolleror microprocessor is used and firmware, software, or microcode can bestored in a tangible or non-transitory storage medium that is associatedwith the device. Any such device may be referred to herein as a“processor” or a “microprocessor.” Such a medium may be a memoryintegrated into the processor, or may be a memory chip that is addressedby the controller to perform control functions. Such firmware, softwareor microcode is executable by the processor and when executed, causesthe processor to perform its control functions. Such firmware orsoftware could also be stored in or on a tangible medium such as anoptical disk or traditional removable or fixed magnetic medium such as adisk drive used to load the firmware or software into an RFID system.

It should be noted that any data and information necessary to supportthe execution of instructions for any embodiment of the disclosure canbe placed in a removable storage medium as well. These could be storedon a disk as well, especially for development purposes or formaintenance and update purposes. Such a storage medium may be accessedeither directly or over a network, including the Internet.

Any of the example embodiments may include additional interpolationstages in the transmitter which for clarity are not shown in thediagrams. Direct conversion and polar modulation radio architectureshave been discussed in this disclosure, but other radio architecturessuch as super-heterodyne or envelope tracking transmitters are alsopossible and do not depart from this disclosure. Although specificembodiments have been illustrated and described herein, those ofordinary skill in the art appreciate that any arrangement which iscalculated to achieve the same purpose may be substituted for thespecific embodiments shown and that the disclosure has otherapplications in other environments. The following claims are in no wayintended to limit the scope of the disclosure to the specificembodiments described herein.

The example embodiments described in this disclosure or alternativeembodiments may be implemented as discrete component RFID readerdesigns, such as using physically separate chips for DACs, ADCs, mixers,amplifiers, mux, couplers, and the like. The waveform memory may beSRAM, DDR, FLASH, or other types of memory internal or external to theDSP processor. The DSP processor may be a digital signal processor suchas a Blackfin® processor from Analog Devices Inc., or it could be a moregeneral purpose microprocessor such as one of the many variants of ARMprocessors, or the DSP processor could be implemented on a fieldprogrammable gate array (FPGA), or on an application specific integratedcircuit (ASIC). The RFID reader implementing the disclosed concepts mayalso be implemented as a system-on-a-chip (SoC), wherein many of thesubsystems such as the DSP, DACs, ADCs, mixers, local oscillators, etc.,are integrated together on a chip. Sometimes multichip SoC solutions areused to ease the manufacturability given the variations in process whichmay be required based on frequency, power, and the like. Any discrete orintegrated form the RFID reader may take which implements the disclosedideas does not depart from the ideas disclosed herein.

The invention claimed is:
 1. A radio frequency (RF) system comprising:an amplitude modulation (AM) transmit signal generator to produce afully modulated AM signal representing a sequence of symbols; an RFamplifier; and an orthogonal offset generator connected to the AMtransmit signal generator and the RF amplifier to introduce a constant,approximately orthogonal offset that is uncorrelated with the sequenceof symbols to shift a trajectory of the fully modulated AM signal awayfrom an origin and create an amplitude and phase modulated offset AM(OAM) transmit signal with a modulation depth reduced by the constant,approximately orthogonal offset to mitigate distortion in the RFamplifier.
 2. The RF system of claim 1 further comprising an RF sourceto produce a carrier wave.
 3. The RF system of claim 2 furthercomprising a digital predistortion block connected after the AM transmitsignal generator and the orthogonal offset generator.
 4. The RF systemof claim 2 further comprising a transmitter nulling offset generator toadd a transmitter nulling offset to the OAM signal.
 5. The RF system ofclaim 2 further comprising a receiver to receive an incoming signal. 6.The RF system of claim 1 further comprising an absolute value functionand an angle function connected to produce the OAM transmit signal as apolar signal.
 7. The RF system of claim 1 wherein the amplitude andphase-modulated OAM transmit signal comprises continuous phasemodulation.
 8. The RF system of claim 1 wherein the OAM transmit signalcomprises an offset phase reversal amplitude shift keying (OPR-ASK)signal including at least one of linear or nonlinear predistortion. 9.The RF system of claim 1 wherein the OAM signal comprises an M-arysignal.
 10. A method comprising: providing an amplitude modulation (AM)transmit waveform representing at least one symbol; and orthogonallyoffsetting the AM transmit waveform by using a constant, approximatelyorthogonal offset that is uncorrelated with the at least one symbol toshift a trajectory away from an origin and produce a modulated offset AM(OAM) transmit signal with a modulation depth reduced by the constant,approximately orthogonal offset to mitigate distortion in an RFamplifier.
 11. The method of claim 10 further comprising scaling the AMtransmit waveform.
 12. The method of claim 10 further comprising addingpredistortion to the modulated OAM signal.
 13. The method of claim 12wherein the predistortion comprises at least one of linear predistortionor nonlinear predistortion.
 14. The method of claim 10 furthercomprising adding a transmitter nulling offset to the modulated OAMtransmit signal.
 15. The method of claim 10 further comprisingconverting the modulated OAM transmit signal to polar coordinates. 16.The method of claim 10 wherein the producing of the AM transmit waveformcomprises generating the AM transmit waveform using Nyquist filtering orquadratic programming.
 17. The method of claim 10 wherein the OAMtransmit signal comprises an M-ary signal.
 18. Apparatus comprising:means for providing an amplitude modulation (AM) transmit waveformrepresenting at least one symbol; and means for orthogonally offsettingthe AM transmit waveform using a controlled constant value that isuncorrelated with the at least one symbol to shift a trajectory awayfrom an origin and produce a modulated offset AM (OAM) transmit signalwith a modulation depth reduced by an approximately orthogonal offsetcorresponding to the controlled constant value signal to mitigatedistortion in an RF amplifier.
 19. The apparatus of claim 18 wherein theOAM transmit signal comprises an offset phase reversal amplitude shiftkeying (OPR-ASK) signal including at least one of linear or nonlinearpredistortion.
 20. The apparatus of claim 18 wherein the OAM signalcomprises an M-ary signal.